NTSC interference rejection filter

ABSTRACT

An electronic, programmable filter is disclosed which selectively removes interference, noise or distortion components from a frequency band without perturbing any of the other signals of the band. An input frequency band such as a television channel spectrum is initially demodulated to baseband and applied to the input of the filter. The baseband spectrum is combined in a complex mixer with a synthesized frequency signal that shifts the spectrum a characteristic amount, in the frequency domain, so as to position an interference component in the region about DC. Once shifted, the frequency components about DC are removed by DC canceler circuit and the resulting spectrum is mixed with a subsequent synthesized frequency signal which shifts the spectrum back to its original representation and baseband. The frequency signals are developed by a programmable frequency synthesizer which a user may program with an intelligence signal that defines the frequency location of an interference signal within the spectrum. Filter blocks may be added or subtracted in order to optimize the filter response for any number of interference components for which rejection is desired.

CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] This application is a continuation of application Ser. No.10/004,515 filed Nov. 2, 2001, which is a continuation of applicationSer. No. 09/685,476 filed Oct. 10, 2000 (now U.S. Pat. No. 6,344,871),which is a continuation of application Ser. No. 09/303,783 filed Apr.30, 1999 (now U.S. Pat. No. 6,219,088), which claimed the benefit ofProvisional Application No. 60/106,938 filed Nov. 3, 1998, thedisclosures of which are incorporated fully herein.

FIELD OF THE INVENTION

[0002] The invention relates generally to television signal transmissionsystems and methods and, more particularly, to a system and method foreliminating the effects of NTSC analog television signal interferencecomponents on digital advanced television (DATV) signals when both aresimultaneously transmitted in the same frequency band.

BACKGROUND OF THE INVENTION

[0003] Recent years have witnessed the establishment of a standard fortransmission of high definition television (HDTV) signals, over bothcable and terrestrial broadcast modes throughout the United States.Although it offers significantly enhanced picture resolution,terrestrial broadcast of HDTV signals is somewhat problematic due to thealmost universal installed base of conventional NTSC broadcast and moreparticularly, reception equipment. The present system provides forsimultaneous transmission (simulcast broadcasting) of HDTV signals andconventional NTSC analog television signals in order to provide highdefinition television services without obsoleting the installed base ofNTSC receivers. Conceptually; program material is encoded into the twodifferent formats (NTSC and HDTV) and simultaneously broadcast overrespective 6 MHz transmission channels. Viewers having conventional NTSCequipment would be able to receive and view NTSC programs by tuning inthe appropriate NTSC channel, while viewers equipped with HDTV equipmentwould be able to receive an HDTV program by tuning their receiver to theappropriate HDTV channel. While conceptually simple, simultaneousbroadcast of NTSC and HDTV signals often results in characteristicportions of an NTSC signal interfering with adjacent channel orco-channel HDTV signals causing degradation to the HDTV signal.

[0004] The cause of this form of signal degradation is well understoodby those familiar with high definition television transmission systemsand is conventionally termed NTSC co-channel interference. Various meanshave been proposed in the art to reduce NTSC co-channel interference incurrent HDTV transmission methodologies, and particularly with respectto vestigial sideband (VSB) HDTV transmissions, which form the basis ofthe HDTV standard in the United States. Certain of these conventionalNTSC interference rejection means are summarized in ATSC standard A/53(1995) ATSC Digital Television Standard. Briefly, the interferencerejection properties of a conventional HDTV system are based on thefrequency location of the principal components of the NTSC co-channelinterfering signal within the 6 MHz television channel.

[0005]FIG. 1 depicts a typical 6 MHz channel spectrum, represented inbaseband in the frequency domain (i.e., symmetric about DC), andillustrated in its characteristic raised cosine form 10 with rootNyquist band edges. NTSC co-channel interference is generally recognizedas caused by the three principal carrier components of an NTSC signal;the video carrier (also termed the luma or luminance carrier), the colorsubcarrier (also termed the chroma or chrominance subcarrier), and theaudio carrier (also termed the aural carrier). In the illustrativechannel spectrum diagram of FIG. 1, the location and approximatemagnitudes of the three principal NTSC components are depicted with thevideo carrier, indicated at V, located approximately 1.25 MHz from thelower channel band edge. The color subcarrier, C, is locatedapproximately 3.58 MHz above the video carrier frequency and the audiocarrier, A, is located approximately 4.5 MHz above the video carrierfrequency (i.e., approximately 0.25 MHz from the upper channel bandedge). As depicted in the Figure, and as well understood in the art,NTSC carrier component interference is of particular concern due to therelatively large amplitudes of the video carrier V and color subcarrierC which characterize and NTSC transmission. Although the audio carrier Ais present at a relatively smaller amplitude, it neverthelesscontributes a significant interference characteristic. Thus, it will beunderstood that NTSC co-channel interference rejection is an importantconsideration in the design of HDTV reception equipment. The carrier andsubcarrier components of an interfering NTSC signal must be removed froman HDTV channel in order to ensure the enhanced quality of an HDTVsignal.

[0006] A conventional approach to NTSC co-channel interference rejectionis based on the frequency location of the principal components of theNTSC co-channel interfering signal within the 6 MHz HDTV channel and theperiodic nulls of a conventional twelve symbol, feed-forward,subtractive, baseband comb filter, disposed conventionally in thedemodulation path of a typical prior art-type VSB receiver.

[0007] Such a conventional baseband comb filter is depicted insemi-schematic block diagram form in FIG. 2 and suitably comprises a 1tap linear feed-forward filter, indicated generally at 12, which can berepresented as in terms of a feed-forward delay stage 13 providing aninverted, delayed, input component to a composite adder 14. Such combfilters are well understood by those having skill in the art and itscomponent parts and principals of operation require no furtherexplanation herein. It will suffice to state that the delay stage 13 isconstructed such that the filter produces an output spectrum havingperiodic spectral nulls equally spaced about 57×f_(H) (896.85 kHz)apart, where f_(H) is equal to the NTSC horizontal line rate. Thus, asshown in FIG. 3, there are 7 periodic nulls occurring within the 6 MHzchannel band, with the NTSC video carrier frequency V fallingapproximately 2.1 kHz below the second null of the comb filter, thecolor subcarrier C falling near the sixth null, and the audio carrier Afalling approximately 13.6 kHz above the seventh null.

[0008] Although the comb filter (12 of FIG. 2) has been generallyadopted by the television transmission and reception industry, itsuffers from certain significant disadvantages that make its universaluse problematic. While providing rejection of steady-state signalslocated at the null frequencies, only the NTSC color subcarrier C iscorrectly placed in the center of the filter's sixth null frequency. Thevideo and audio carriers V and A occur at frequencies that are offsetfrom their respective filter null positions. This prevents the NTSCvideo and audio carrier signals from being completely canceled by thefilter. In addition to incomplete rejection of the NTSC interferencecomponents, the filter also has the effect of modifying data signalswhich occur at the location of the periodic nulls throughout the 6 MHzHDTV channel. Although the modified data signal can be recovered andsomewhat properly decoded by a trellis decoder, the complexity of such adecoder is substantially increased, particularly when it is recognizedthat the number of slicing levels, comprising the decision loop, willnecessarily be increased from 8 to 15 (a consequence of the partialresponse process characterizing the system).

[0009] Moreover, the effects of channel band noise may be significantlyincreased by the filter. This results, in part, by the reproduction ofnoise appearing on the input line in the filter's delay stage 13, suchthat the filter output contains an accumulation of a noise componentthrough the delay stage 13 and a noise component contained in theoriginal signal. As mentioned above, the conventional comb filter isgenerally effective in rejecting steady-state signal components. Mostforms of noise, however, are random in frequency, phase and amplitude.Many situations will necessarily occur when noise components areadditive, and the resulting noise product may significantly interferewith desired signals, thereby substantially degrading the quality of anHDTV signal.

[0010] Accordingly, there remains a need in the art of HDTV transmissionand reception system design, for a more effective system and method ofreducing the effects of NTSC co-channel interference. Such a systemshould be able to selectively and precisely remove interfering NTSCcarrier component signals without substantial effect on the remainder ofthe channel spectrum (i.e., on user significant data). Further, thesystem should be able to process input channel data and remove unwantedinterference components without introducing extraneous noise and withoutskewing the channel, thereby maintaining the original simplicity of thedemodulator block.

SUMMARY OF THE INVENTION

[0011] One aspect of the present invention is to provide a system andmethod which enables the removal of unwanted signal components, such asNTSC co-channel interference components, from composite channelinformation without introducing extraneous noise and without perturbingany of the remaining signal components of the channel.

[0012] In one particular aspect of the invention, an electronicprogrammable filter is configured to selectively reject NTSC co-channelinterference components from a composite input signal representing anHDTV channel. Channel information has been demodulated to baseband priorto being introduced to the filter input. The programmable filtersuitably comprises at least one programmable digital frequencysynthesizer which defines a signal oscillating at a specificcharacteristic frequency corresponding to a frequency characteristic ofan NTSC co-channel interference component whose removal is desired. Thefrequency signal from the digital frequency synthesizer is combined withthe input channel information in a complex mixer, thereby shifting thechannel spectrum by an amount equal to the synthesized frequency signal,thereby positioning the NTSC co-channel interference component in aregion symmetrical about DC. A DC cancellation circuit removes a narrowband of frequencies about DC, consequently removing the NTSC co-channelinterference component from the channel spectrum.

[0013] Following DC cancellation, the channel spectrum is subsequentlyshifted (upconverted) back to its original baseband representation bycombining the shifted spectrum with an additional frequency signaldeveloped by an additional digital frequency synthesizer.

[0014] The digital frequency synthesizer is programmable in thatparticular shift frequencies are defined by a user and communicated tothe synthesizer circuit through an intelligence signal; the digitalfrequency synthesizer faithfully producing a frequency signaloscillating at the user commanded frequency.

[0015] In a further aspect of the invention, the programmable filterincludes a multiplicity of filter blocks, with each filter blockconfigured to shift an input spectrum by a characteristic frequencyamount and then to cancel the frequency components in a narrow bandabout DC. Each filter block further includes a digital frequencysynthesizer operating to develop a characteristic frequency signal eachfrequency signal corresponding to a displacement metric of the variousNTSC co-channel interference components. The filter operates to shiftthe input spectrum by a first characteristic frequency thereby shiftinga first NTSC co-channel interference component to DC, canceling the DCcomponent and then shifting the resulting spectrum in accordance with asecond characteristic frequency synthesized by a second digitalfrequency synthesizer to thereby shift the spectrum such that a secondNTSC co-channel interference component is positioned at DC. This secondcomponent is removed by a second DC cancellation circuit and the processis repeated, as necessary, for additional interference components whichare present in the input spectrum at definable frequencies.

[0016] Following cancellation of all unwanted interference components,the input spectrum may be returned to baseband by a further filterelement operatively responsive to a further digital frequencysynthesizer which produces a frequency signal having a characteristicfrequency which corresponds to the algebraic sum of all prior shiftfrequencies.

[0017] More specifically, in one embodiment of the invention eachdigital frequency synthesizer might output a sinusoidal waveform havingboth in-phase and quadrature phase components, with the sinusoidalwaveform oscillating at the characteristic frequency a. Further, eachrespective one of the DC cancellation circuits may be tunable so as toexhibit a cancellation bandwidth of from about 5 Hz to about 2 kHz.According to this aspect of the invention modularity of the programmablefilter allows filter elements to be added or removed at need so as toconfigure the filter to reject any number of interference componentsfrom a frequency band, no matter how introduced, so long as thecharacteristic frequencies of the interference components can bedetermined with reasonable precision.

BRIEF DESCRIPTION OF THE DRAWINGS

[0018] These and other features, aspects and advantages of the presentinvention will be more fully understood when considered with respect tothe following detailed description, appended claims and accompanyingdrawings, wherein:

[0019]FIG. 1 is a semi-schematic representation of a conventionalchannel spectrum, illustrated in the frequency domain and in baseband,showing the positions and relative magnitudes of NTSC co-channelinterference components according to the prior art;

[0020]FIG. 2 is a semi-schematic block level diagram of an NTSCinterference rejection filter according to the prior art;

[0021]FIG. 3 is a semi-schematic representation of the output frequencyresponse spectrum of the prior art NTSC interference rejection filter ofFIG. 2, illustrating the positions and spacing of the output nulls withrespect to the NTSC co-channel interference components;

[0022]FIG. 4 is a semi-schematic block level diagram of a generalizedembodiment of the present invention;

[0023]FIG. 5 is a semi-schematic block level diagram of an electronic,programmable, NTSC interference rejection filter constructed accordingto practice of principles of the present invention;

[0024]FIG. 6a-e are a set of semi-schematic frequency diagramsillustrating the operation of the NTSC interference rejection filter ofFIG. 5 in canceling the luminance carrier signal;

[0025]FIG. 7 is a semi-schematic block diagram of a common baseband DCcancellation circuit;

[0026]FIG. 8 semi-schematic block diagram of a QDDFS and complexmultiplier combination in accordance with the principles of theinvention.

DETAILED DESCRIPTION OF THE INVENTION

[0027] In general terms, the present invention may be aptly described asa selectively programmable electronic filter capable of attenuatingspecific, user defined, frequencies within a selected input frequencyspectrum, such as a 6 MHz television channel spectrum, withoutintroducing spurious and unnecessary poles or zeros to the channel andthus, degrading the image quality of an HDTV signal.

[0028]FIG. 4 is a semi-schematic block diagram which depicts theinvention in its generalized form. Conceptually, the invention operatesupon a selected input frequency spectrum by shifting the spectrum withinthe frequency domain by a selected amount and then removing a narrowband of frequencies disposed about a particular reference frequency. Inthe case where it is advantageous to remove a number of undesirablefrequency components disposed throughout the input frequency spectrum,the invention replicates the frequency shift and narrow band removalelements a suitable number of times to accommodate all of theundesirable frequency components encompassed by the input frequencyspectrum. Thus, as illustrated in FIG. 4, the electronic filteraccording to the invention processes an input signal in a firstfrequency shift block 30, thereby shifting the input signal by a firstpreselected amount, to position a first interference or distortioncomponent with respect to a defined reference frequency. Subsequent tothe first frequency shift 30 the signal is directed to a firstcancellation block 31 which selectively cancels a narrow band offrequencies in an area related to the reference frequency, therebydefining a first intermediate signal related to the input signal byhaving been frequency shifted by a first selected amount and by having anarrow band of frequencies, corresponding to a first interference ordistortion component, removed therefrom.

[0029] As shown in FIG. 4, the first intermediate signal is thendirected to a second frequency shift block 32 where the intermediatesignal is shifted, in the frequency domain, by a second predeterminedamount. Subsequently, the signal is directed to a second cancellationblock 33 where a second narrow band of frequencies corresponding to asecond undesirable interference or distortion component is removed fromthe signal, thereby defining a second intermediate signal. The processmay be repeated as many times as desired, with each intermediate signalfrequency shifted and a narrow band of frequencies canceled therefrom inorder to define third, fourth, fifth, etc., intermediate frequenciesuntil all of the interference or distortion components identified forremoval have been canceled from the input signal. Following frequencyshifting and cancellation, the final intermediate signal is directed toa frequency reshift block 34 where the signal is returned to itsoriginal spectrum. Accordingly, the invention may be thought of asencompassing a plurality of steps of frequency shifting and frequencycanceling an input signal, with each set of frequency shift andfrequency cancellation steps defining a corresponding intermediatesignal with a particular interference or distortion component removedtherefrom. If the signal is to be reshifted to its original spectrumrepresentation, the number of shifting steps will necessarily be greaterthan the number of cancellation steps.

[0030] Each frequency shifter includes a programmable frequency sourcethat defines the preselected amount by which the input signal, orintermediate signal, is shifted and which is related to a particularinterference or distortion component whose removal is desired. In oneaspect of the invention, the programmable frequency source functions toshift the input, or intermediate, signal such that the interference ordistortion component is shifted to DC, whence the interference ordistortion components are removed by canceling a narrow band offrequencies disposed about DC. Cancellation bandwidth is necessarily afunction of the characteristics of the interference or distortioncomponent whose removal is desired. If the interference or distortioncomponent is represented by a sharp, narrow band signal, the frequencycancellation block need only cancel those frequencies disposed a few10's of Hertz to either side of DC. Where the interference or distortioncomponent is spread over a relatively wider band, the bandwidth of thefrequency canceller may be varied accordingly such that frequenciesdisposed a few kilohertz to either side of DC are included in thecancellation bandwidth.

[0031] In other words, the present invention attenuates specific, userdefined frequencies within an input signal by frequency shifting andsubsequently DC canceling the input signal until all desiredinterference or distortion components are removed. The inventionfunctions equally well with regard to a single interference ordistortion component or a multiplicity of interference or distortioncomponents. No matter the number of interference or distortioncomponents desired for removal, all that is required is that for eachcomponent, the signal be frequency shifted to bring the component intorelationship with a reference frequency and that frequenciescorresponding to the interference or distortion component aresubsequently canceled from the signal. The signal may or may not bereshifted to its original spectrum depending on the design requirementsof downstream components. If the signal is reshifted to its originalspectrum, the signal is characterized by only having interference ordistortion components removed therefrom.

[0032] A particular, exemplary embodiment of the novel selectivelyprogrammable electronic filter, is depicted in the semi-schematic blockdiagram of FIG. 5, in complex signal form, and is operationallyconfigured to function as an NTSC co-channel interference rejectionfilter. In the illustrated embodiment, an NTSC interference rejectionfilter, generally indicated at 40, may be thought of as comprising asequentially disposed bank of generally similar filter elements, eachconstructed and functioning in accord with principles of the invention.When configured to reject NTSC interference components, the filter isimplemented in three stages 42, 44 and 46 each of which are configuredto remove one of the three NTSC co-channel interference components (thevideo carrier, color subcarrier and audio carrier) from a typical 6 MHzHDTV channel spectrum.

[0033] A complex-valued input signal, identified as u(t) andrepresenting a recovered complex-valued baseband channel spectrum,including NTSC co-channel interference components, is received by thefilter from, for example, a preceding front-end channel tuner block (notshown) of an exemplary HDTV receiver. It should be noted, however, thatneither the input frequency spectrum nor the filter's position in aparticular application need be specified with any particularity in orderto practice the principles of the invention. The exemplary applicationof the filter embodiment of FIG. 5, the signal conventions, names andwhether the signals are complex or real-valued, are employed solely forillustrative purposes and not as limitations to the scope of theinvention.

[0034] The complex-valued input u(t), is received at the input of thefirst filter block 42, within which u(t) is directed to and provides oneof the complex inputs (I & Q) to a dual port complex multiplier stage50. In the complex multiplier stage 50, the complex input signal u(t) ismodulated (combined), in a process conventionally termed downconversionwhich will be described in greater detail below, with the complex-valuedoutput of a high-performance direct digital frequency synthesizer(termed a DDFS herein) stage 52. In accord with the invention, the DDFSstage 52 synthesizes phase-coherent periodic signals at specific,programmable, user defined frequencies, denoted in the illustration ofFIG. 5 generally as e-^(1ωt), which are used to modulate (downconvert)the complex input signal u(t) in the complex multiplier 50. At thisjuncture, it should be mentioned that the notational form of thecharacteristic output signal of the direct digital frequency synthesizer(DDFS), e-^(iωt), assumes the signal is complex-valued. The signal mayalso be viewed as resolved into real-valued components, a sine(ωt)component and a cos(ωt) component without disturbing the example. In theexample of FIG. 5, the direct digital frequency synthesizer (DDFS)characteristic output signal is expressed as a complex-valuedexponential purely for the sake of convenience.

[0035] With further reference to FIG. 5, the first filter block 42comprises a DDFS 52 which produces a complex-valued signal at a firstcharacteristic, user defined frequency, e-^(iω1t), where ω1 is the firstuser definable frequency index. Combining the input signal u(t) with thefirst characteristic signal e-^(iω1t) synthesized by the DDFS 52, in thecomplex multiplier 50, has the effect of shifting the HDTV channelfrequency spectrum represented by u(t), up or down, by a specific amountequal to the characteristic frequency ω1 of the DDFS 52, such that thecharacteristic frequency ω1 of the 6 MHz channel band is made tocoincide with DC in a frequency domain baseband representation. Thus,the complex multiplier 50 and DDFS 52, in combination, function tofrequency shift, or downconvert, the composite frequency elements of thechannel spectrum until the channel frequency corresponding to thecharacteristic frequency of the first DDFS, ω, is shifted to DC.

[0036] Following signal conversion, the downconverted complex signal,now represented in FIG. 5 as X(t), is directed to the input of abaseband DC canceler block 54 where frequency components in the regionabout DC are removed from the channel spectrum, i.e., canceled. Inaccordance with the present invention, the baseband DC canceler 54 iscapable of suppressing signals in the DC region, with a programmablecancellation bandwidth ranging from about 6 Hz to about 2 KHz, with eachof the cancellation ranges being symmetric about the DC axis inbaseband. Thus, a range of frequencies, having a selectivelyprogrammable bandwidth, symmetric about wl, is removed from the channelspectrum.

[0037] The utility of this approach will be more clearly understood ifthe characteristic frequency ω1 of the conversion signal synthesized bythe first DDFS block 52 is recognized as corresponding to the videocarrier component's frequency spacing from the channel band edge, i.e.,1.25 MHz. As depicted more clearly in the semi-schematic frequencyspectrum diagrams of FIGS. 6(a) to 5(e), the complex multiplier 50 ofthe first filter block 42 mixes the baseband spectrum of FIG. 6(a) withthe DDFS output e-iω1t, thereby shifting (downconverting) the basebandspectrum a characteristic amount wl, and positioning the NTSC videocarrier component coincident with DC, as depicted in FIG. 6(b). A notchis introduced, symmetric about DC, by operation of the baseband DCcanceler block 54, thereby removing the NTSC video carrier interferencecomponent signal from the 6 MHz channel spectrum.

[0038] Returning now to FIG. 5, once the video carrier interferencecomponent signal is removed, the resulting complex signal, representedas X(t), is output from the first baseband DC canceler 54 and directedto the input of the second, programmable, electronic filter stage 44where it is applied to one input of a second, dual port, complexmultiplier 56. In a manner similar to that described in connection withthe first filter block, a second direct digital frequency synthesizerblock 58 synthesizes a second complex-valued signal, having a secondcharacteristic frequency, e-^(iω2t), which is combined with the outputof the first filter block X(t) in the second complex multiplier 56. Thesecond complex multiplier 56 functions to again shift the frequencyspectrum by a characteristic amount equal to the characteristicfrequency ω2 of the signal synthesized by the second DDFS block 58, suchthat frequencies once displaced from DC by ω2 now are coincident withDC. Once the spectrum has been shifted by the desired amount, thefrequencies. in a range about DC are once again removed (canceled) by asecond baseband DC canceler block 60.

[0039] The operation of the second programmable electronic filter block44 and its resultant effect on the channel spectrum may be furtherunderstood with reference to the semi-schematic frequency spectrumdiagrams of FIGS. 6(b) and 5(c). In FIG. 6(b) it can be seen that theNTSC color subcarrier is located a distance of ω2 from the videocarrier, in the frequency domain. After the video carrier has beenremoved, FIG. 6(c) illustrates the resulting channel frequencydistribution after the signal has been shifted once again, i.e.,downconverted by a characteristic amount equal to the characteristicfrequency w2 of signal synthesized by the second DDFS 58 of the secondprogrammable electronic filter block 44. It should be noted that thesecond characteristic frequency ω2 is equal to approximately 3.58 MHzand represents the frequency difference between the NTSC colorsubcarrier interference component and the NTSC video carrierinterference, at least as they are presently defined. Thus, it will beunderstood that the second programmable electronic filter bank 44functions to shift the channel spectrum such that the color subcarrierinterference component is made to coincide with baseband DC and then toapply a DC cancellation to a narrow band of frequencies, symmetricalabout DC, such that the color subcarrier interference component isremoved from the channel spectrum.

[0040] In similar fashion, the third programmable electronic filter bank46 receives the complex-valued input signal Y(t) output from thebaseband DC canceler 60 of the second programmable electronic filterbank 44 and directs Y(t) to the input of a third, dual port, complexmultiplier 64. In a manner similar to that described in connection withthe first and second filter blocks, a third direct digital frequencysynthesizer block 62 synthesizes a third complex-valued signal,e-^(iω3t), having a third characteristic frequency ω3. This third signalis once again combined with the output of the second filter block Y(t)in the third complex multiplier 64, and the resultant complex-valuedsignal, now denoted Z(t), is directed to the input of a third basebandDC canceler 66. In the same fashion as the first and second programmableelectronic filter blocks 42 and 44, the third programmable electronicfilter block 46 functions to shift the channel spectrum a third amountequal to the characteristic frequency ω3 of the downconversion signaldeveloped in the third DDFS block 62 as illustrated in the spectrumrepresentation of FIG. 6(d).

[0041] As illustrated in FIG. 6(d), the characteristic frequency ω3developed by the third DDFS block 62 corresponds to the frequencyspacing between the NTSC color subcarrier interference component and theNTSC audio carrier interference component. As is well understood in theart, the NTSC audio carrier component position of the spectrum may beviewed in a number of ways. Conventionally, the audio carrier isvariously described as being spaced 0.25 MHz from the top end of thechannel spectrum, 4.5 MHz from (above) the video carrier, or about 0.92MHz from (above) the color subcarrier signal. In the context of theillustrated embodiment, particularly with regard to the illustration ofFIG. 6(d), the third characteristic frequency ω3 developed by the thirdDDFS block 62 is equal to approximately 0.92 MHz, i.e., the frequencyseparation between the color subcarrier interference component removedin the second filter block 44 and the position of the audio carrierinterference component in the frequency domain.

[0042] Once the three NTSC interference components are removed from theHDTV channel spectrum, it should be recognized that the resultantchannel spectrum Z(t) is displaced in frequency from its initialsymmetry about DC, by an amount equal to the algebraic sum of thefrequency components of the signals synthesized by the various DDFScircuits. Z(t) accordingly must be translated (upconverted) back tou(t), i.e., baseband, prior to further down-stream processing such ascarrier recovery, equalization, and the like. Accordingly, after NTSCinterference component removal, the HDTV channel spectrum is upconvertedin an upconversion stage 68, suitably comprising an additional complexmultiplier 70 and yet a further direct digital frequency synthesizerblock 72. In a manner substantially similar to the DDFS blocks of theprior downconversion and DC cancellation stages, the DDFS block 72 ofthe upconversion stage 70 defines a third complex-valued signale^(i(ω1+ω2+ω3)t) which has a characteristic frequency representing thealgebraic sum of the characteristic frequencies of the DDFS blocks 52,58 and 62 of the prior downconversion and DC cancellation stages 42, 44and 46, respectively. This characteristic synthesized frequency signalis combined with Z(t) in complex multiplier 70 to remove the frequencydisplacement effects of the downconversion and DC canceler stages,thereby, as depicted in the spectrum diagrams of FIGS. 6(d) and 5(e),shifting the channel spectrum back to its initial baseband position,i.e., symmetric about DC.

[0043] Having reference now to FIG. 6(e), it will be evident that theoriginal input frequency spectrum u(t) has been reproduced with a highdegree of fidelity and with relatively little spurious data signal loss.In the case of a complex HDTV channel, the signal output by the filterin accord with the invention, will be characterized by very narrow-bandfrequency notches. Each frequency notch is, in turn, characterized by auser programmable bandwidth in the range of from about 6 Hz to about 2KHz, with each notch corresponding to and coinciding with only thespecific, characteristic frequencies of the video, color and audiocarrier components of an NTSC co-channel interference signal.

[0044] In other words, the programmable electronic filter 40, in accordwith the illustrated embodiment, can be thought of as a combination ofsequential modular elements, each configured to adaptively translate acomplex channel by a specific desired frequency and then cancel thechannel components in the DC region. Multiple elements allow multiplefrequency components to be removed without perturbing the remainingspectrum. Filtration thus occurs only at the specified frequencies.

[0045] Noise amplification in the cancellation regions is no longerproblematic, since the noise response characteristics in these regionsdepend on the characteristics of a DC cancellation circuit. Thesecircuits are well understood by those having skill in the art andrequire no further elaboration herein. However, for completeness, anexemplary DC cancellation circuit, suitable for incorporation in theprogrammable filter of the invention, is illustrated in semi-schematicblock diagram form in FIG. 7.

[0046] Briefly, the DC cancellation circuit of FIG. 7 includes a delayelement 80, represented as an inverse z transform block, whose output isfeedback coupled to a summing circuit 82, where it is negativelycombined with the output of a gain stage 84. The gain stage isconstructed to exhibit a gain (K) which is typically less than unity inthe exemplary embodiment.

[0047] The output of the delay element 80 is also negatively combinedwith an input signal (In) in an input summing circuit 86, the output ofwhich defines the output of the cancellation circuit, as well as theinput of the gain stage 84.

[0048] In operation, the exemplary DC cancellation circuit of FIG. 7functions to remove a band of frequencies from the input signal (In),with the frequencies and the band characteristics defined by the filterelement design values. In the present case, the center frequency is DCand the filter cancellation bandwidth may be modulated by varying thedesign parameters of the inverse z transform block 80 and the gain stage84.

[0049] A particular implementation of a DDFS (more specifically aquadrature direct digital frequency synthesizer or QDDFS), and complexmixer combination, suitable for implementation in an NTSC interferencerejection filter according to the present invention, is depicted insemi-schematic block diagram form in FIG. 8. Conventionally, the DDFS,indicated generally at 100, is termed a QDDFS since it comprises a pairof operational signal paths operating in quadrature with each other, soas to develop sine and cosine signals having specific phase argumentsand at particular frequencies. Sine and cosine signals are able to bedeveloped at multiple frequencies, with any particular frequency ofinterest being generated in operative response to a multi-bit frequencycontrol word (FCW). The frequency control word is input to a phaseaccumulator, or alternatively an integrator, 102 which generates thephase argument of the sine/cosine function. In conventional fashion, thebinary structure of the frequency control word identifies theparticular. frequency at which the complimentary sine/cosine waveformsare to be generated.

[0050] A sine/cosine function generator is suitably implemented, in theillustrated embodiment, as dual sine/cosine ROM look-up tables, each ofwhich contains a digital representation of a respective sine/cosinewaveform. Specifically, a representative periodic waveform (a completesine wave, for example) is resolved into 2¹² discrete samples with eachsample defining a discrete point of the waveform. Each of the 2¹²samples are stored, as 10-bit characters, in respective 2¹² addressablelocations in each respective ROM look-up table (sine and cosine). Asaddressable locations are sequentially accessed and output, the samplevalues reconstruct sine or cosine waveforms.

[0051] Frequency of the output waveform is determined by the argument ofthe frequency control word which defines the sampling spacing of samplestaken from the digitized sine/cosine waveform. By way of example, if thesampling spacing (address spacing) were zero, the same address would beaccessed, the same 10-bit sample would be returned for every outputclock, and the output frequency would be zero. Depending on the inputphase argument, the output signal level might also be zero, or mightinclude a positive or negative DC component. If the address samplespacing were one, each and every 10-bit sample word would be output, onesample word at each clock. Thus, the output waveform would exhibit afrequency equal to f_(c)/4096, where f_(c) is the clock frequency. Asaddress sample spacing. increases, i.e., coarser waveform definition,the output frequency necessarily increases. Further, those having skillin the art will understand that the maximum output frequency of a DDFSconstructed in accordance with the embodiment of FIG. 8, willnecessarily be limited by the Nyquist sampling criterion, sometimesexpressed as about one half (½) the chosen clock frequency f_(c).

[0052] Once a sine and cosine waveform, having the appropriate phase andfrequency characteristics, are synthesized, the sine and cosinewaveforms are mixed with the in-phase and quadrature-phase signals I andQ, which are the real-valued equivalents to the complex frequencyspectrum (u(t), X(t), Y(t), or Z(t)) to be modulated, in a complex mixer106. The construction of the complex mixer 106 may be conventional inform and functions to combine the two sets of signals in accord withcomplex multiplication principles, to achieve an equivalentcomplex-valued product; in the case of the illustrated embodiment, themixer 106 shifts the complex-valued frequency spectrum represented by Iand Q, by an amount e^(-jωt), where ω represents the frequency of theoutput sine/cosine waveforms sin(ωt) and cos(ωt), to define Iout andQout, where:

Iout=[I sin(ωt)−Q cos(ωt)]; and

Qout=i [I cos(ωt)+Q sin(ωt)]

[0053] Various alternative QDDFS architectures will become apparent toone having skill in the art, when one considers the quarter-wavesymmetry of a sine wave and the n/2 phase relationship between sine andcosine waveforms. An alternative DDFS architecture might be constructedto take advantage of the quarter wave symmetry of a sine wave in orderto reduce ROM storage requirements. Only sine samples from 0 to n/2 maybe stored in the ROM look-up table and a quadrant designator bit, suchas a second most significant bit developed by the integrator, might usedto determine the waveform quadrant, thereby synthesizing a sine wavefrom 0 to n. The most significant bit might then used as a sign bit inorder to complete the synthesis of the sine wave from 0 to 2n. In thecase of a cosine waveform, its 0 crossings are advanced by n/2 withrespect to those of a sine waveform. Thus, the MSB from the integratormight be EXORed with the second MSB in order to generate the sign bitand synthesize a complete cosine wave from 0 to 2n.

[0054] For designs where quadrature outputs are desired, theabove-described system would store both sine and cosine samples, from 0to n/2, in respective ROM look-up tables. Alternatively, a systemdesigner could take advantage of the eighth wave symmetry of a sine andcosine wave form, since sine samples from 0 to n/4 are the same ascosine samples from n/4 to n/2. Similarly, cosine samples from 0 to n/4are the same as sine samples from n/4 to n/2. Thus, one need only storesine and cosine samples from 0 to n/4. The third MSB from the integratormight be used to select between the samples, with the third MSB EXORedwith the second MSB in order to produce the selection signal. This lastis necessary such that the select signal is phase aligned with theeighth wave symmetry axis plane of the sine and cosine waveform.

[0055] Although various embodiments of a DDFS circuit have beendescribed, these descriptions are for illustrative purposes only and arenot intended as limiting the invention to any particular adaptation of afrequency synthesis circuit. The actual form of the DDFS implementationis immaterial to practice of the invention, so long as the DDFS iscapable of synthesizing periodic waveforms having user definablefrequency characteristics.

[0056] Up to this point, the programmable, electronic filter accordingto the invention, has been described in connection with its usefulnessin terrestrial mode HDTV television and reception systems, andparticularly in connection with NTSC co-channel interference rejection.Notwithstanding the foregoing, it will be immediately recognized thatthe programmable, electronic filter according to the invention hasparticular utility in a substantial number of varying and differentdigital communication system applications. One exemplary alternativeapplication is the use of the programmable electronic filter in CATVreception equipment as a means to provide selective and efficientrejection of spurious, non-random noise or interference signals, i.e.,distortion, pertinent to CATV-type transmission media and CATV-typesystem architectures.

[0057] One of the advantages of broadband transmission media is itsability to allow a substantial expansion of the number of transmissionchannels available for television signals, including HDTV cablecasts.Additional channels are available at higher and higher frequencies and,at least for the immediate future, will be populated with both NTSCstandard signals and HDTV signals. Given the large numbers of signalchannels present in a modern composite channel, it should not bedifficult to understand that the channel's very capacity allows a largenumber of distortion sources to coexist with a desired HDTV channel, forexample. Although the type of adjacent channel distortion, which givesrise to NTSC interference in terrestrial broadcast modes, is not presentin transmission line architectures, other means of introducing NTSCinterference signals into an HDTV channel are indeed present.Specifically, channel impairment might be caused by a variety of wellunderstood distortion sources, such as “signal to second harmonic”distortion and “signal to third harmonic” distortion effects which, whensummed over the frequency spectra of all of the channels on atransmission line, can give rise to substantial, frequency specific,noise spikes. Video carrier, color subcarrier and audio carrier signalswhich are generated by a particular NTSC transmitter down-band from, forexample, an HDTV channel, are also replicated by the NTSC transmitter inmultiple harmonic sidebands. Although a carrier or subcarrier componentin. a harmonic sideband is attenuated, the sideband is neverthelessaccessible to higher and lower order harmonics produced by other NTSCtransmitters that are further down-band or up-band. Fortunately, sincethe amplitude characteristics of these sidebands approach a negligiblelevel, the farther away they are from the fundamental and since the datacomposites of these sidebands are random with respect to one another,composite data signals from up-band or down-band transmitters would notimpair the performance of a selected-HDTV channel. However, the NTSCcomponents are quite frequency specific and share the same phaserelationships. Thus, even HDCATV reception equipment must include somemeans for removing these accumulated, frequency specific, distortioncomponents.

[0058] Extensive tests and evaluations have been performed on variousCATV systems to determine the properties of signal distortions, i.e.,composite second order (CSO) and composite triple beat (CTB), as afunction of the number and frequencies of television channels comprisinga CATV system. Distortion tends to significantly increase in directproportion to the number of channels in the system and also tends toincrease in direct proportion to channel frequency. However, since themost prevalent distortion sources are frequency specific, thesedistortions can be characterized with respect to frequency and removedfrom an input HDTV signal in accordance with the present invention.

[0059] Indeed, the programmable electronic filter of the invention hasapplication to any communication system no matter what the channel mediaand no matter what the source of any frequency specific component deemedto be impairing the channel. Because of its modular nature, theimpairment source might comprise a single signal, in which case only asingle filter stage will be necessary, or might comprise multiplesignals, in which case each signal will spawn a corresponding filterstage. All that is required is that an interference source bedeterminable and occur at a specific, known frequency, such that theinterference source frequency can form the basis of a corresponding DDFSsynthesized signal.

[0060] A programmable electronic filter has been described in thecontext of illustrated embodiments directed to NTSC co-channelinterference rejection. Filter functions to shift a frequency spectrumby a pre-determined amount to thereby center an interference sourceabout DC, prior to canceling a narrow band of frequencies about DC inorder to remove the interference source from a frequency spectrum.Subsequent interference components are removed by subsequently shiftingthe frequency spectrum in any desired direction for any desiredfrequency distance. After all of the interference components areremoved, frequency spectrum is returned to baseband or to any otherfrequency location, by a final frequency shift process.

[0061] While the invention has been described in terms of operating oncomplex-valued signals, it will be evident to one having skill in theart that complex-valued signals may be easily resolved into theirreal-value analogs and that the electronic filter according to theinvention is equally capable of functioning in real-value terms. It willthus be recognized by those skilled in the art that variousmodifications may be made to the illustrated and other embodiments ofthe invention described above, without departing from the broadinventive scope thereof. It will be understood therefore that theinvention is not limited to the particular embodiments or arrangementsdisclosed but is rather intended to include any changes, adaptations ormodifications which are within the scope and spirit of the invention asdefined by the appended claims.

What is claimed is:
 1. A method for rejecting at least one particularinterference frequency component from an input frequency spectrum, themethod comprising: shifting the input spectrum in the frequency domaindown by a first selected amount, thereby positioning.the particularinterference frequency component about DC, as a result of the shift;canceling the shifted input spectrum in the region about DC in order toremove the particular interference frequency component from the inputspectrum; and shifting the input spectrum in the frequency domain up bya second selected amount.